View APU3037_1108985.PDF datasheet online --- IC-ON-LINE

Datasheet File OCR Text:

Technology Licensed from International Rectifier
APU3037 / APU3037A
8-PIN SYNCHRONOUS PWM CONTROLLER DESCRIPTION
The APU3037 controller IC is designed to provide a low cost synchronous Buck regulator for on-board DC to DC converter applications. With the migration of today's ASIC products requiring low supply voltages such as 1.8V and lower, together with currents in excess of 3A, traditional linear regulators are simply too lossy to be used when input supply is 5V or even in some cases with 3.3V input supply. The APU3037 together with dual N-channel MOSFETs such as AP60T03, provide a low cost solution for such applications. This device features an internal 200KHz oscillator (400KHz for "A" version), under-voltage lockout for both Vcc and Vc supplies, an external programmable soft-start function as well as output under-voltage detection that latches off the device when an output short is detected.
FEATURES
Synchronous Controller in 8-Pin Package Operating with single 5V or 12V supply voltage Internal 200KHz Oscillator (400KHz for APU3037A) Soft-Start Function Fixed Frequency Voltage Mode 500mA Peak Output Drive Capability Protects the output when control FET is shorted RoHS Compliant
APPLICATIONS
DDR memory source sink Vtt application Low cost on-board DC to DC such as 5V to 3.3V, 2.5V or 1.8V Graphic Card Hard Disk Drive
TYPICAL APPLICATION
5V 12V C3 0.1uF C4 1uF C2 10TPB100M, 100uF, 55mV L1 1uH C1 47uF
Vcc
Vc HDrv Q1 AP60T03GH 1N4148 LDrv Q2 AP60T03GH R3 249, 1% Gnd R5 1.24K, 1% L2 1.5V/5A 5.6uH, 5.3A C7 2x 6TPC150M, 150uF, 40mV
SS/SD C8 0.1uF
U1 APU3037
C9 2200pF R4 24k
Comp
Fb
Figure 1 - Typical application of APU3037 or APU3037A.
PACKAGE ORDER INFORMATION
TA (C) 0 To 70 0 To 70 0 To 70 0 To 70 DEVICE APU3037O APU3037M/MP APU3037AO APU3037AM/AMP PACKAGE FREQUENCY 8-Pin Plastic TSSOP (O) 200KHz 8-Pin Plastic SOIC NB (M/MP) 200KHz 8-Pin Plastic TSSOP (O) 400KHz 8-Pin Plastic SOIC NB (M/MP) 400KHz
Data and specifications subject to change without notice.
200507075-1/18
APU3037 / APU3037A
ABSOLUTE MAXIMUM RATINGS
Vcc Supply Voltage .................................................. Vc Supply Voltage ...................................................... Storage Temperature Range ...................................... Operating Junction Temperature Range ..................... 25V 30V (not rated for inductive load) -65C To 150C 0C To 125C
CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device.
PACKAGE INFORMATION
8-PIN PLASTIC TSSOP (O) 8-PIN PLASTIC SOIC (M/MP)
Fb 1 Vcc 2 LDrv 3 Gnd 4
8 SS/SD 7 Comp 6 Vc 5 HDrv
Fb 1 Vcc 2 LDrv 3 Gnd 4
8 SS/SD 7 Comp 6 Vc 5 HDrv
JA=124C/W
JA=160C/W without exposed pad (M) JA=80C/W with exposed pad (MP)
ELECTRICAL SPECIFICATIONS
Unless otherwise specified, these specifications apply over Vcc=5V, Vc=12V and TA=0 to 70C. Typical values refer to TA=25C. Low duty cycle pulse testing is used which keeps junction and case temperatures equal to the ambient temperature. PARAMETER Reference Voltage Fb Voltage Fb Voltage Line Regulation UVLO UVLO Threshold - Vcc UVLO Hysteresis - Vcc UVLO Threshold - Vc UVLO Hysteresis - Vc UVLO Threshold - Fb UVLO Hysteresis - Fb Supply Current Vcc Dynamic Supply Current Vc Dynamic Supply Current Vcc Static Supply Current Vc Static Supply Current Soft-Start Section Charge Current SYM
VFB LREG
TEST CONDITION APU3037 APU3037A 5MIN 1.225 0.784
TYP 1.250 0.800 0.2 4.2 0.25 3.3 0.2 0.6 0.4 0.1 5 7 3.3 1 -20
MAX 1.275 0.816 0.35 4.4 3.5 0.8 0.5
UNITS
V % V V V V V V
UVLO Vcc Supply Ramping Up UVLO Vc Supply Ramping Up UVLO Fb Fb Ramping Down (APU3037)
4.0 3.1 0.4 0.3
(APU3037A)
Dyn Icc Dyn Ic ICCQ ICQ SSIB
Freq=200KHz, CL=1500pF Freq=200KHz, CL=1500pF SS=0V SS=0V SS=0V
2 2 1 0.5 -10
8 10 6 4.5 -30
mA mA mA mA mA
2/18
APU3037 / APU3037A
PARAMETER Error Amp Fb Voltage Input Bias Current Fb Voltage Input Bias Current Transconductance Oscillator Frequency Ramp-Amplitude Voltage Output Drivers Rise Time Fall Time Dead Band Time Max Duty Cycle Min Duty Cycle SYM
IFB1 IFB2
TEST CONDITION SS=3V, Fb=1V SS=0V, Fb=1V
MIN
TYP -0.1 -64 600 200 400 1.25 50 50 150 90 0
MAX
UNITS mA mA mmho KHz V ns ns ns % %
gm
Freq VRAMP Tr Tf TDB TON TOFF
450 APU3037 APU3037A 180 360 1.225
750 220 440 1.275 100 100 250 95
CL=1500pF CL=1500pF Fb=1V, Freq=200KHz Fb=1.5V 50 85 0
PIN DESCRIPTIONS
PIN# 1 PIN SYMBOL PIN DESCRIPTION Fb This pin is connected directly to the output of the switching regulator via resistor divider to provide feedback to the Error amplifier. Vcc This pin provides biasing for the internal blocks of the IC as well as power for the low side driver. A minimum of 1mF, high frequency capacitor must be connected from this pin to ground to provide peak drive current capability. Output driver for the synchronous power MOSFET. This pin serves as the ground pin and must be connected directly to the ground plane. A high frequency capacitor (0.1 to 1mF) must be connected from V5 and V12 pins to this pin for noise free operation. Output driver for the high side power MOSFET. Connect a diode, such as BAT54 or 1N4148, from this pin to ground for the application when the inductor current goes negative (Source/ Sink), soft-start at no load and for the fast load transient from full load to no load. This pin is connected to a voltage that must be at least 4V higher than the bus voltage of the switcher (assuming 5V threshold MOSFET) and powers the high side output driver. A minimum of 1mF, high frequency capacitor must be connected from this pin to ground to provide peak drive current capability. Compensation pin of the error amplifier. An external resistor and capacitor network is typically connected from this pin to ground to provide loop compensation. This pin provides soft-start for the switching regulator. An internal current source charges an external capacitor that is connected from this pin to ground which ramps up the output of the switching regulator, preventing it from overshooting as well as limiting the input current. The converter can be shutdown by pulling this pin below 0.5V.
2
3 4
LDrv Gnd
5
HDrv
6
Vc
7
Comp
8
SS / SD
3/18
APU3037 / APU3037A
BLOCK DIAGRAM
Vcc 0.2V Bias Generator 3V 1.25V
4.0V 3V Vc 20uA 3.5V SS/SD 8 POR 1.25V Error Comp 25K Error Amp R 25K Fb 1 FbLo Comp 0.5V Comp 7 POR Reset Dom Q 64uA Max Oscillator Ct S 0.2V
POR
6 Vc
5 HDrv
2 Vcc
3 LDrv
4 Gnd
Figure 2 - Simplified block diagram of the APU3037.
THEORY OF OPERATION
Introduction The APU3037 is a fixed frequency, voltage mode synchronous controller and consists of a precision reference voltage, an error amplifier, an internal oscillator, a PWM comparator, 0.5A peak gate driver, soft-start and shutdown circuits (see Block Diagram). The output voltage of the synchronous converter is set and controlled by the output of the error amplifier; this is the amplified error signal from the sensed output voltage and the reference voltage. This voltage is compared to a fixed frequency linear sawtooth ramp and generates fixed frequency pulses of variable duty-cycle, which drives the two N-channel external MOSFETs.The timing of the IC is provided through an internal oscillator circuit which uses on-chip capacitor to set the oscillation frequency to 200 KHz (400 KHz for "A" version). Soft-Start The APU3037 has a programmable soft-start to control the output voltage rise and limit the current surge at the start-up. To ensure correct start-up, the soft-start sequence initiates when the Vc and Vcc rise above their threshold (3.3V and 4.2V respectively) and generates the Power On Reset (POR) signal. Soft-start function operates by sourcing an internal current to charge an external capacitor to about 3V. Initially, the soft-start function clamps the E/A's output of the PWM converter. As the charging voltage of the external capacitor ramps up, the PWM signals increase from zero to the point the feedback loop takes control. Short-Circuit Protection The outputs are protected against the short-circuit. The APU3037 protects the circuit for shorted output by sensing the output voltage (through the external resistor divider). The APU3037 shuts down the PWM signals, when the Fb voltage drops below 0.6V (0.4V for APU3037A). The APU3037 also protects the output from over-voltaging when the control FET is shorted. This is done by turning on the sync FET with the maximum duty cycle. Under-Voltage Lockout The under-voltage lockout circuit assures that the MOSFET driver outputs remain in the off state whenever the supply voltage drops below set parameters. Lockout occurs if Vc and Vcc fall below 3.3V and 4.2V respectively. Normal operation resumes once Vc and Vcc rise above the set values.
4/18
APU3037 / APU3037A
APPLICATION INFORMATION
Design Example: The following example is a typical application for APU3037, the schematic is Figure 18 on page 14. VIN = 5V VOUT = 3.3V IOUT = 4A DVOUT = 100mV fS = 200KHz Output Voltage Programming Output voltage is programmed by reference voltage and external voltage divider. The Fb pin is the inverting input of the error amplifier, which is internally referenced to 1.25V (0.8V for APU3037A). The divider is ratioed to provide 1.25V at the Fb pin when the output is at its desired value. The output voltage is defined by using the following equation: VOUT = VREF 3 Soft-Start Programming The soft-start timing can be programmed by selecting the soft start capacitance value. The start up time of the converter can be calculated by using:
tSTART = 753Css (ms)
---(2)
Where: CSS is the soft-start capacitor (mF) For a start-up time of 7.5ms, the soft-start capacitor will be 0.1mF. Choose a ceramic capacitor at 0.1mF. Shutdown The converter can be shutdown by pulling the soft-start pin below 0.5V. The control MOSFET turns off and the synchronous MOSFET turns on during shutdown. Boost Supply Vc To drive the high-side switch it is necessary to supply a gate voltage at least 4V greater than the bus voltage. This is achieved by using a charge pump configuration as shown in Figure 18. The capacitor is charged up to approximately twice the bus voltage. A capacitor in the range of 0.1mF to 1mF is generally adequate for most applications. In application, when a separate voltage source is available the boost circuit can be avoided as shown in Figure 1. Input Capacitor Selection The input filter capacitor should be based on how much ripple the supply can tolerate on the DC input line. The larger capacitor, the less ripple expected but consider should be taken for the higher surge current during the power-up. The APU3037 provides the soft-start function which controls and limits the current surge. The value of the input capacitor can be calculated by the following formula: CIN = IIN 3 Dt DV ---(3)
(
R6 1+ R5
)
---(1)
When an external resistor divider is connected to the output as shown in Figure 3.
VOUT APU3037
Fb R5 R6
Figure 3 - Typical application of the APU3037 for programming the output voltage. Equation (1) can be rewritten as: R6 = R5 3
( VOUT - 1) VREF
Choose R5 = 1KV This will result to R6 = 1.65KV If the high value feedback resistors are used, the input bias current of the Fb pin could cause a slight increase in output voltage. The output voltage set point can be more accurate by using precision resistor.
Where: CIN is the input capacitance (mF) IIN is the input current (A) Dt is the turn on time of the high-side switch (ms) DV is the allowable peak to peak voltage ripple (V)
5/18
APU3037 / APU3037A
Assuming the following: DV = 1%(VIN), Efficiency(h) = 90% 1 fS VO 3 IO IIN = h 3 VIN Dt = D 3 Dt = 3.3ms IIN = 2.93A Di 1 VOUT ; Dt = D3 ;D= Dt fS VIN VOUT L = (VIN - VOUT)3 ---(5) VIN3Di3fS Where: VIN = Maximum Input Voltage VOUT = Output Voltage i = Inductor Ripple Current fS = Switching Frequency t = Turn On Time D = Duty Cycle VIN - VOUT = L3 If Di = 20%(IO), then the output inductor will be: L = 7mH The Toko D124C series provides a range of inductors in different values, low profile suitable for large currents, 10mH, 4.2A is a good choice for this application. This will result to a ripple approximately 14% of output current. Power MOSFET Selection The APU3037 uses two N-Channel MOSFETs. The selections criteria to meet power transfer requirements is based on maximum drain-source voltage (VDSS), gatesource drive voltage (VGS), maximum output current, Onresistance RDS(ON) and thermal management. The MOSFET must have a maximum operating voltage (VDSS) exceeding the maximum input voltage (VIN). The gate drive requirement is almost the same for both MOSFETs. Logic-level transistor can be used and caution should be taken with devices at very low VGS to prevent undesired turn-on of the complementary MOSFET, which results a shoot-through current. The total power dissipation for MOSFETs includes conduction and switching losses. For the Buck converter the average inductor current is equal to the DC load current. The conduction loss is defined as: PCOND (Upper Switch) = ILOAD 3 RDS(ON) 3 D 3 q PCOND (Lower Switch) = ILOAD 3 RDS(ON) 3 (1 - D) 3 q q = RDS(ON) Temperature Dependency The RDS(ON) temperature dependency should be considered for the worst case operation. This is typically given in the MOSFET data sheet. Ensure that the conduction losses and switching losses do not exceed the package ratings or violate the overall thermal budget.
2 2
By using equation (3), CIN = 193.3mF For higher efficiency, low ESR capacitor is recommended. Choose two 100mF capacitors. The Sanyo TPB series PosCap capacitor 100mF, 10V with 55mV ESR is a good choice. Output Capacitor Selection The criteria to select the output capacitor is normally based on the value of the Effective Series Resistance (ESR). In general, the output capacitor must have low enough ESR to meet output ripple and load transient requirements, yet have high enough ESR to satisfy stability requirements. The ESR of the output capacitor is calculated by the following relationship: ESR [ DVO DIO ---(4)
Where: DVO = Output Voltage Ripple DIO = Output Current DVO=100mV and DIO=4A Results to ESR=25mV The Sanyo TPC series, PosCap capacitor is a good choice. The 6TPC150M 150mF, 6.3V has an ESR 40mV. Selecting two of these capacitors in parallel, results to an ESR of 20mV which achieves our low ESR goal. The capacitor value must be high enough to absorb the inductor's ripple current. The larger the value of capacitor, the lower will be the output ripple voltage. Inductor Selection The inductor is selected based on output power, operating frequency and efficiency requirements. Low inductor value causes large ripple current, resulting in the smaller size, but poor efficiency and high output noise. Generally, the selection of inductor value can be reduced to desired maximum ripple current in the inductor (i). The optimum point is usually found between 20% and 50% ripple of the output current. For the buck converter, the inductor value for desired operating ripple current can be determined using the following relation:
6/18
APU3037 / APU3037A
For this design, AP60T03GH is a good choice. The device provides low on-resistance in a TO-252 package. The AP60T03GH has the following data: VDSS = 30V ID = 4.5A RDS(ON) = 0.012V The total conduction losses will be:
PCON(TOTAL)=PCON(Upper Switch)+PCON(Lower Switch)
These values are taken under a certain condition test. For more detail please refer to the AP60T03GH data sheet. By using equation (6), we can calculate the switching losses. PSW = 0.127W Feedback Compensation The APU3037 is a voltage mode controller; the control loop is a single voltage feedback path including error amplifier and error comparator. To achieve fast transient response and accurate output regulation, a compensation circuit is necessary. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency and adequate phase margin (greater than 458). The output LC filter introduces a double pole, -40dB/ decade gain slope above its corner resonant frequency, and a total phase lag of 1808 (see Figure 5). The Resonant frequency of the LC filter expressed as follows: FLC = 1 2p3 LO3CO ---(7)
PCON(TOTAL) = ILOAD 3 RDS(ON) 3 q q = 1.5 according to the AP60T03GH data sheet for 1508C junction temperature PCON(TOTAL) = 0.288W The switching loss is more difficult to calculate, even though the switching transition is well understood. The reason is the effect of the parasitic components and switching times during the switching procedures such as turn-on / turnoff delays and rise and fall times. With a linear approximation, the total switching loss can be expressed as: VDS(OFF) tr + tf 3 ILOAD PSW = 3 ---(6) 2 T Where: VDS(OFF) = Drain to Source Voltage at off time tr = Rise Time tf = Fall Time T = Switching Period ILOAD = Load Current The switching time waveform is shown in figure 4.
VDS 90%
2
Figure 5 shows gain and phase of the LC filter. Since we already have 1808 phase shift just from the output filter, the system risks being unstable.
Gain 0dB -40dB/decade Phase 08
-1808 FLC Frequency
FLC
Frequency
Figure 5 - Gain and phase of LC filter. The APU3037's error amplifier is a differential-input transconductance amplifier. The output is available for DC gain control or AC phase compensation.
tr td(OFF) tf
10% VGS td(ON)
Figure 4 - Switching time waveforms. From AP60T03GH data sheet we obtain:
tr = 57.5ns tf = 6.4ns
The E/A can be compensated with or without the use of local feedback. When operated without local feedback the transconductance properties of the E/A become evident and can be used to cancel one of the output filter poles. This will be accomplished with a series RC circuit from Comp pin to ground as shown in Figure 6.
7/18
APU3037 / APU3037A
Note that this method requires that the output capacitor should have enough ESR to satisfy stability requirements. In general the output capacitor's ESR generates a zero typically at 5KHz to 50KHz which is essential for an acceptable phase margin. The ESR zero of the output capacitor expressed as follows: 1 FESR = ---(8) 2p 3 ESR 3 Co
VOUT R6 Fb
Where: VIN = Maximum Input Voltage VOSC = Oscillator Ramp Voltage Fo = Crossover Frequency FESR = Zero Frequency of the Output Capacitor FLC = Resonant Frequency of the Output Filter R5 and R6 = Resistor Dividers for Output Voltage Programming gm = Error Amplifier Transconductance For: VIN = 5V VOSC = 1.25V Fo = 30KHz FESR = 26.52KHz FLC = 2.9KHz R5 = 1K R6 = 1.65K gm = 600mmho This results to R4=104.4KV. Choose R4=105KV To cancel one of the LC filter poles, place the zero before the LC filter resonant frequency pole: FZ 75%FLC
R5 VREF
Gain(dB)
E/A
Comp V e C9 R4
H(s) dB
FZ
Frequency
Figure 6 - Compensation network without local feedback and its asymptotic gain plot. The transfer function (Ve / VOUT) is given by: H(s) = gm 3
FZ 0.75 3
1 2p LO 3 CO
---(13)
(
R5 R6 + R5
sR4C9 ) 3 1 +sC9
---(9)
For: Lo = 10mH Co = 300mF FZ = 2.17KHz R4 = 86.6KV Using equations (11) and (13) to calculate C9, we get:
The (s) indicates that the transfer function varies as a function of frequency. This configuration introduces a gain and zero, expressed by: R5 |H(s)| = gm3 3 R4 ---(10) R63R5 FZ = 1 2p3R43C9 ---(11)
C9 = 698pF Choose C9 = 680pF One more capacitor is sometimes added in parallel with C9 and R4. This introduces one more pole which is mainly used to supress the switching noise. The additional pole is given by: 1 FP = C9 3 CPOLE 2p 3 R4 3 C9 + CPOLE The pole sets to one half of switching frequency which results in the capacitor CPOLE: CPOLE = 1 1 p3R43fS C9 fS for FP << 2 1 p3R43fS
The gain is determined by the voltage divider and E/A's transconductance gain. First select the desired zero-crossover frequency (Fo): Fo > FESR and FO [ (1/5 ~ 1/10)3 fS Use the following equation to calculate R4: R4 = 1 VOSC Fo3FESR R5 + R6 3 3 3 gm VIN FLC2 R5 ---(12)
8/18
APU3037 / APU3037A
For a general solution for unconditionally stability for any type of output capacitors, in a wide range of ESR values we should implement local feedback with a compensation network. The typically used compensation network for voltage-mode controller is shown in Figure 7.
ZIN C10 R8 R6 Fb R5 VREF VOUT C12 R7 C11 Zf
FP1 = 0 FP2 = FP3 = 1 2p3R83C10 1 C123C11 2p3R73 C12+C11 1 2p3R73C11
(
)
1 2p3R73C12
FZ1 =
1 1 FZ2 = 2p3C103(R6 + R8) 2p3C103R6 Cross Over Frequency: VIN 1 FO = R73C103 3 VOSC 2p3Lo3Co Where: VIN = Maximum Input Voltage VOSC = Oscillator Ramp Voltage Lo = Output Inductor Co = Total Output Capacitors
E/A
Comp
V e
---(15)
Gain(dB)
H(s) dB
FZ1
FZ2
FP2
FP3
Frequency
Figure 7 - Compensation network with local feedback and its asymptotic gain plot. In such configuration, the transfer function is given by: 1 - gmZf Ve = VOUT 1 + gmZIN The error amplifier gain is independent of the transconductance under the following condition:
The stability requirement will be satisfied by placing the poles and zeros of the compensation network according to following design rules. The consideration has been taken to satisfy condition (14) regarding transconductance error amplifier. 1) Select the crossover frequency: Fo < FESR and Fo [ (1/10 ~ 1/6)3 fS 2) Select R7, so that R7 >> 2
gmZf >> 1
and
gmZIN >>1
gm
---(14)
By replacing ZIN and Zf according to Figure 7, the transformer function can be expressed as: H(s)= 1 3 sR6(C12+C11)
[
(1+sR7C11)3[1+sC10(R6+R8)] C123C11 1+sR7 3(1+sR8C10) C12+C11
3) Place first zero before LC's resonant frequency pole. FZ1 75% FLC 1 C11 = 2p 3 FZ1 3 R7 4) Place third pole at the half of the switching frequency. FP3 = fS 2 1 2p 3 R7 3 FP3
(
)]
As known, transconductance amplifier has high impedance (current source) output, therefore, consider should be taken when loading the E/A output. It may exceed its source/sink output current capability, so that the amplifier will not be able to swing its output voltage over the necessary range. The compensation network has three poles and two zeros and they are expressed as follows:
C12 =
C12 > 50pF If not, change R7 selection. 5) Place R7 in (15) and calculate C10: C10 [ 2p 3 Lo 3 Fo 3 Co VOSC 3 R7 VIN
9/18
APU3037 / APU3037A
6) Place second pole at the ESR zero. FP2 = FESR 1 R8 = 2p 3 C10 3 FP2 Check if R8 > 1 Start to place the power components, make all the connection in the top layer with wide, copper filled areas. The inductor, output capacitor and the MOSFET should be close to each other as possible. This helps to reduce the EMI radiated by the power traces due to the high switching currents through them. Place input capacitor directly to the drain of the high-side MOSFET, to reduce the ESR replace the single input capacitor with two parallel units. The feedback part of the system should be kept away from the inductor and other noise sources, and be placed close to the IC. In multilayer PCB use one layer as power ground plane and have a control circuit ground (analog ground), to which all signals are referenced. The goal is to localize the high current path to a separate loop that does not interfere with the more sensitive analog control function. These two grounds must be connected together on the PC board layout at a single point. Figure 8 shows a suggested layout for the critical components, based on the schematic on page 14.
PGnd
C1 C2A, B
gm
If R8 is too small, increase R7 and start from step 2. 7) Place second zero around the resonant frequency. FZ2 = FLC 1 R6 = - R8 2p 3 C10 3 FZ2 8) Use equation (1) to calculate R5. R5 = VREF 3 R6 VOUT - VREF
These design rules will give a crossover frequency approximately one-tenth of the switching frequency. The higher the band width, the potentially faster the load transient speed. The gain margin will be large enough to provide high DC-regulation accuracy (typically -5dB to 12dB). The phase margin should be greater than 458 for overall stability. IC Quiescent Power Dissipation Power dissipation for IC controller is a function of applied voltage, gate driver loads and switching frequency. The IC's maximum power dissipation occurs when the IC operating with single 12V supply voltage (Vcc=12V and Vc24V) at 400KHz switching frequency and maximum gate loads. Figures 9 and 10 show voltage vs. current, when the gate drivers loaded with 470pF, 1150pF and 1540pF capacitors. The IC's power dissipation results to an excessive temperature rise. This should be considered when using APU3037A for such application. Layout Consideration The layout is very important when designing high frequency switching converters. Layout will affect noise pickup and can cause a good design to perform with less than expected results.
PGnd PGnd
C7A, B
L1 Vin
8 7 6 5
Vout
L2 Q1
1 2 3 4
DDDC 3215
5
4
C4 6
PGnd
R4 C9 7
U1 3 APU3037
2 C3 R5 1 R6
Single Point Analog Gnd Connect to Power Ground plane Analog Gnd
C8 8
Analog Gnd
Figure 8 - Suggested layout. (Topside shown only)
10/18
APU3037 / APU3037A
TYPICAL PERFORMANCE CHARACTERISTICS
APU3037A Vcc vs. Icc
@470PF, 1150PF and 1540PF Gate Load
TA = 258C
14 12 10 8 6 4 2 0 0 2 4 6 Vcc (V) 8
CLOAD =1540pF CLOAD =1150pF
Icc (mA)
CLOAD =470pF
10
12
14
Figure 9 - Vcc vs. Icc
APU3037A Vc vs. Ic
@470PF, 1150PF and 1540PF Gate Load
TA = 258C
30 25 Ic (ma) 20 15 10 5 0 0 2 4 6 8 10 12 14 16 18 20 22 24 26 Vc (V)
Figure 10 - Vc vs. Ic
CLOAD =470pF CLOAD =1540pF CLOAD =1150pF
11/18
APU3037 / APU3037A
TYPICAL PERFORMANCE CHARACTERISTICS
APU3037 Output Voltage
1.3 240
APU3037 Output Frequency
230
1.28 220 Max
Max
210 1.26 Kilo Hertz 1.24 190
Volts
200
Min
180 1.22
Min
170
1.2 -40C 0C Output Voltage +50C Spec Max. +100C Spec Min. +150C
160 -40C 0C +50C +100C Spec Max. Spec Min. +150C
Oscillation Frequency
Figure 11 - Output Voltage
Figure 12 - Output Frequency
APU3037 Maximum Duty Cycle
92.0%
90.0%
88.0% Percent Duty Cycle
86.0%
84.0%
82.0%
80.0% -40C -25C 0C +25C +50C +75C +100C +125C +150C
Max Duty Cycle
Figure 13 - Maximum Duty Cycle
12/18
APU3037 / APU3037A
TYPICAL PERFORMANCE CHARACTERISTICS
APU3037A Output Voltage
820 460
APU3037A Output Frequency
Max 440 810 Max
420
800 400
milli Volts
790
Kilo Hertz
380
Min
780
360
Min
340
770 320
760 -40C -25C 0C +25C +50C Spec Max. +75C +100C +150C
300 -40C -25C 0C +25C +50C +75C +100C +150C
Output Voltage
Spec Min.
Oscillation Frequency
Spec Max.
Spec Min.
Figure 14 - Output Voltage
APU3037 / APU3037A Transconductance ( GM )
Figure 15 - Output Frequency
APU3037 / APU3037A Rise Time / Fall Time CL = 1500pF
1000 50 900 45 800 40 700 35 600 nano Seconds micro Mho's
30
500
25
400
20
300
15
200
10
100
5
0 -40C -25C 0C +25C +50C Negative load GM +75C +100C
0 -40C -25C 0C +25C Rise Time +50C Fall time +75C +100C
Positive load GM
Figure 16 - Transconductance
Figure 17 - Rise Time and Fall Time
13/18
APU3037 / APU3037A TYPICAL APPLICATION
Single Supply 5V Input
5V D1 1N4148 D3 1N4148 D2 1N4148 C4 1uF C5 0.1uF L1 1uH C2 2x 10TPB100ML, 100uF, 55mV C1 47uF Tantalum
C3 0.1uF
Vcc
Vc HDrv Q1 AP60T03GH 1N4148 L2 3.3V @ 4A C7 2x 6TPC150M, 150uF, 40mV
SS/SD C8 0.1uF
U1 APU3037
LDrv
744311470 4.7uH Q2 AP60T03GH
Comp C9 680pF Gnd R4 105K R5 1K, 1% Fb
R6 1.65K, 1%
Figure 18 - Typical application of APU3037 in an on-board DC-DC converter using a single 5V supply.
14/18
APU3037 / APU3037A
TYPICAL APPLICATION
Dual Supply, 5V Bus and 12V Bias Input
5V L1 12V 1uH C2 10TPB100M, 100uF, 55m V, 1.5A rms C1 47uF 1.8V/1A APU1206-18
Vcc V c HDrv SS/SD U1 APU3037
C4 0.1uF
C1 1uF
Q1 AP60T03GH 1N4148 Q2 AP60T03GH R1
C3 47uF L2 3.5uH @ 2.5A C6 6TPB150M, 150uF, 55mV 2.5V/2A
C7 0.1uF
LDrv
C8 2200pF R2 14K
Cm op Fb
1K, 1%
Gnd
R3 1K 1%
L2 3.5uH @ 2.5A 5.7uH @ 2.5A
C6 6TPB150M, 150uF, 55m V (Qty 2) 6TPB150M, 150uF, 55m V (Qty 1)
C9 10TPB100M, 100uF, 55m V, 1.5A rms C10 0.1uF C11 1uF
Vcc
V c HDrv
Q3 AP60T03GH 1N4148 Q4 AP60T03GH R4
L3 3.4uH @ 2A C12 6TPB150M, 150uF, 55mV
3.3V/1.8A
SS/SD
C13 0.1uF
U2 APU3037
LDrv
Cm op
C14 2200pF R5 14K
Fb
1.65K, 1%
Gnd
R6 1K, 1%
Figure 19 - Typical application of APU3037 or APU3037A in an on-board DC-DC converter providing the Core, GTL+, and Clock supplies for the Pentium II microprocessor.
15/18
APU3037 / APU3037A
TYPICAL APPLICATION
1.8V to 7.5V / 0.5A Boost Converter
Vpwr (1.5V Min)
L1 1uH
Vc/Vcc
C1 2x 68uF
C5 1uF
D1 1N5817
VOUT (7.5V / 0.5A)
R1 20K R2 10K Q1 2N2222 R3 20K Q2 2N2222 C10 100pF C4 0.01uF R4 25K C5 0.1uF
C9 2x 47uF Q3 AP60T03GH
SS/SD Comp Vc U1
HDrv
APU3037
Fb Vcc LDrv Gnd
C8 1uF
Gnd
R5 1K, 1% R6 5K, 1%
Figure 20 - Typical application of APU3037 as a boost converter.
16/18
APU3037 / APU3037A
DEMO-BOARD APPLICATION
5V or 12V to 3.3V @ 10A
L1
VIN 5V or 12V
1uH C1 33uF 16V D4 LL4148 C2A 47uF 16V C2B 47uF 16V D2 LL4148
D1 LL4148
Gnd
C3 1uF
C19 1uF C6 0.1uF L2
Vcc
C8 1uF
Vc HDrv
Q1 AP60T03GH
SS/SD
C5 0.1uF
U1 APU3037
LDrv
1N4148 Q2 AP60T03GH C7 470pF R4 4.7V
3.2uH
VOUT 3.3V @ 10A
C9B 150uF 6.3V C9C 150uF 6.3V C13 1uF
Comp Gnd
C4 2200pF R3 20K R6
Gnd
Fb
1.65K R5 1K
Figure 21 - Demo-board application of APU3037.
Application Parts List Ref Desig Q1 Q2 U1 D1, D2, D4 L1 L2 C1 C2A, C2B C9B, C9C C5, C6 C3 C4 C7 C8, C13, C19 R3 R4 R5 R6 Description MOSFET MOSFET Controller Diode Inductor Inductor Capacitor, Tantalum Capacitor, Poscap Capacitor, Poscap Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Resistor Resistor Resistor Resistor Value Qty Part# 30V, 12mV, 45A 1 AP60T03GH 30V, 12mV, 45A 1 AP60T03GH Synchronous PWM 1 APU3037 Fast Switching 3 LL4148 1mH, 10A 1 7445601 3.2mH, 12A 1 7443550320 33mF, 16V 1 ECS-T1CD336R 47mF, 16V, 70mV 2 16TPB47M 150mF, 6.3V, 40mV 2 6TPC150M 0.1mF, Y5V, 25V 2 ECJ-2VF1E104Z 1mF, X7R, 25V 1 ECJ-3YB1E105K 2200pF, X7R, 50V 1 ECJ-2VB1H222K 470pF, X7R 1 ECJ-2VB2D471K 1mF, Y5V, 16V 3 ECJ-2VF1C105Z 20K, 5% 1 4.7V, 5% 1 1K, 1% 1 1.65K, 1% 1 Manuf APEC APEC APEC WE WE
17/18
APU3037 / APU3037A
DEMO-BOARD WAVEFORMS
APU3037
VIN=5.0V, V OUT =3.3V
100
VIN
Efficiency (%)
90
VOUT
80
70 0 1 2 3 4 5 6 7 8 9 10 11
Output Current (A)
Figure 22 - Efficiency for APU3037 Evaluation Board.
Figure 23 - Start-up time @ IOUT=5A.
Vss
APU3037 APU3037
VOUT
IOUT = 5V
Figure 24 - Shutdown the output by pulling down the soft-start.
Figure 25 - 3.3V output voltage ripple @ IOUT=5A.
APU3037
APU3037
2A
4A
0A
0A
Figure 26 - Transient response @ IOUT = 0 to 2A.
Figure 27 - Transient response @ IOUT = 0 to 4A.
18/18

▲Up To Search▲

Price & Availability of APU3037

	To Download APU3037 Datasheet File
If you can't view the Datasheet, Please click here to try to view without PDF Reader .